An integrated circuit often needs a low power supply voltage to avoid heat-dissipation problems. Low supply voltage will be even more necessary in the future as integrated circuits become increasingly more dense. It thus becomes desirable that a differential amplifier used as an input stage in an op amp have rail-to-rail input capability. That is, the output signal of the amplifier should be representative of the differential input signal as its common-mode voltage travels the full range of the power supply voltage. As used here to describe the relationship between two signals, "representative" means that their amplitudes have a substantially one-to-one (typically linear) relationship as long as the amplitudes are not too great.
Referring to the drawings, FIG. 1 illustrates a general arrangement for a prior art differential amplifier that can achieve rail-to-rail input capability at moderately low power supply voltage. See U.S. Pat. No. 4,555,673. Also see U.S. Pat. Nos. 4,463,319 and 4,532,479.
The amplifier in FIG. 1 employs a pair of complementary differential input portions 10 and 12 to amplify a circuit input signal differentially supplied between input terminals T1 and T2 as input voltages V.sub.I1 and V.sub.I2. Differential portion 10 consists of NPN transistors Q1 and Q2 whose bases receive the circuit input signal at input points P1 and P2 connected to terminals T1 and T2. The emitters of transistors Q1 and Q2 are connected together at a node N.sub.A to receive a first operating current I.sub.A. Differential portion 12 is formed with PNP transistors Q3 and Q4 whose bases receive the circuit input signal at input points P3 and P4 connected to terminals T1 and T2. The emitters of transistors Q3 and Q4 are connected together at a node N.sub.B to receive a second operating current I.sub.B. A main current supply 14 connected between sources for a high supply voltage V.sub.HH and a low supply voltage V.sub.LL provides currents I.sub.A and I.sub.B in opposite flow directions.
Portion 10 supplies amplified internal currents I.sub.D1 and I.sub.D2 from the Q1 and Q2 collectors. Portion 12 similarly provides amplified internal currents I.sub.D3 and I.sub.D4 from the Q3 and Q4 collectors. A summing circuit 16 connected between the V.sub.HH and V.sub.LL supplies suitably combines currents I.sub.D1 -ID.sub.4 to generate a pair of complementary output currents I.sub.O and I.sub.O.
In looking at amplifier operation, it is convenient to define several terms. Let ".DELTA.V" and "V.sub.CM " respectively represent the differential voltage V.sub.I1 -V.sub.I2 and the common-mode voltage (V.sub.I1 + V.sub.I2)/2 of the circuit input signal. Let "V.sub.PS " represent the power supply voltage V.sub.HH - V.sub.LL.
The prior art device in FIG. 1 can be characterized by a pair of threshold voltages V.sub.TA and V.sub.TB that may vary with V.sub.PS. FIG. 2 shows a general example of how V.sub.TA and V.sub.TB may appear. FIG. 3 shows the resulting operational regions for the amplifier. The horizontal axis in FIG. 3 represents the condition where V.sub.CM equals V.sub.LL. Line 18 (at 45.degree.) represents the condition where V.sub.CM equals V.sub.HH. The region between the horizontal axis and line 18 thereby encompasses the rail-to-rail operational range for the amplifier.
Input portion 10 is operatively conductive (turned on) when V.sub.CM is sufficiently high. More specifically: EQU V.sub.CM -V.sub.LL .gtoreq.V.sub.TA (1)
Eq. (1) basically corresponds to the portion of the V.sub.PS range between line 20A and line 18 in FIG. 3. The conductive region for portion 10 does, however, extend slightly above line 18 to a point where V.sub.CM is several tenths of a volt more than V.sub.HH. When portion 10 is turned on, it amplifies the voltage difference between points P1 and P2 by splitting current I.sub.A between currents I.sub.D1 and I.sub.D2 at values whose difference is representative of .DELTA.V.
Input portion 12 is operatively conductive when V.sub.CM is sufficiently low. In particular: EQU V.sub.HH -V.sub.CM .gtoreq.V.sub.TB (2)
Eq. (2) basically corresponds to the portion of the V.sub.PS range between the horizontal axis and line 20B in FIG. 3. In addition, the conductive range for portion 10 extends slightly below the horizontal axis to a point where V.sub.CM is several tenths of a volt less than V.sub.LL. hen portion 12 is turned on, it amplifies the voltage difference between points P3 and P4 by splitting current I.sub.B between currents I.sub.D3 and I.sub.D4 at values whose difference is representative of .DELTA.V.
FIG. 3 indicates that portion 10 is turned off in the space between the horizontal axis and line 20A--i.e., when V.sub.CM -V.sub.LL is less than V.sub.TA. Likewise, FIG. 3 shows portion 12 as being turned off in the area between lines 20B and 18. This corresponds to the condition in which V.sub.HH -V.sub.CM is less than V.sub.TB. In actuality, each of portions 10 and 12 usually switches between on and off over a spacing on the order of 100 millivolts. Lines 20A and 20B are thus idealizations of narrow voltage regions.
When V.sub.PS is reduced, the amount of V.sub.CM space available for the non-conductive regions for portions 10 and 12 decreases. The two non-conductive regions start to overlap when V.sub.PS reaches a level V.sub.PSO. If V.sub.PS drops below V.sub.PSO, V.sub.CM passes through a "dead zone" indicated by thick line 22 in which neither of portions 10 and 12 is operatively conductive. V.sub.PSO is thus the value of the minimum V.sub.PS level at which the differential amplifier in FIG. 1 can achieve rail-to-rail input capability.
Thresholds V.sub.TA and V.sub.TB have respective minimum values V.sub.MA and V.sub.MB. See lines 24 and 26 in FIG. 3. In known embodiments of the amplifier in FIG. 1, V.sub.PSO is approximately equal to V.sub.MA + V.sub.MB.
The numerical value of V.sub.PSO depends on the base-emitter voltages of transistors Q1-Q4 and on the internal construction of current supply 14. FIGS. 4a and 4b depict two ways, both described in U.S. Pat. No. 4,555,673, for implementing supply 14. FIGS. 5a and 5b graphically show the specific idealized operational regions for the amplifier of FIG. 1 as implemented with the main current supplies shown respectively in FIGS. 4a and 4b.
Starting with the embodiment of FIG. 4a, supply 14 consists simply of a pair of current sources S.sub.L and S.sub.H. Current source S.sub.L supplies current I.sub.A at a constant value I.sub.L. Current source S.sub.H provides current I.sub.B at a constant value I.sub.H. For current sources S.sub.L and S.sub.H to be conductive, the voltage across each of them must be at least equal to a minimum level V.sub.SAT.
V.sub.CM exceeds the voltage at node N.sub.A by 1V.sub.BE when portion 10 is turned on. V.sub.BE is the magnitude of the standard voltage across the base-emitter junction of a bipolar transistor when it is just turned on. The voltage at node N.sub.B similarly exceeds V.sub.CM by 1V.sub.BE when portion 12 is turned on. In view of this, each of threshold minimums V.sub.MA and V.sub.MB equals V.sub.BE + V.sub.SAT. V.sub.PSO thereby equals 2V.sub.BE + 2V.sub.SAT.
At a standard collector-emitter current of several microamperes, V.sub.BE is approximately 0.6 volt. V.sub.SAT can be as low as 0.1 volt. Using these values for V.sub.BE and V.sub.SAT (in the present computation and in all the additional ones below), V.sub.PSO is approximately 1.4 volts for the differential amplifier of FIGS. 1 and 4a.
A disadvantage of using the current supply in FIG. 4a is that the transconductance of the amplifier changes by a factor of approximately 2 whenever either of portions 10 and 12 turns on or off. The variation in transconductance makes it difficult to optimize the frequency compensation for the amplifier when it is used in an op am with negative feedback. The implementation shown in FIG. 4b overcomes this problem.
When the circuitry of FIG. 4b is used, supply 14 is formed with current source S.sub.H, a PNP steering transistor QP whose base receives a reference voltage V.sub.RP, and a current mirror 28 that supplies current I.sub.A at a value largely equal to the current I.sub.QP through transistor QP. Current steering through transistor QP enables the sum of I.sub.A and I.sub.B to approximately equal I.sub.H. As a result, the transconductance is nearly constant as V.sub.CM varies across the entire V.sub.PS range.
The voltage across current mirror 28 between node N.sub.A and the V.sub.LL supply must be at least 1V.sub.SAT for mirror 28 to be conductive. Under optimum conditions, the current steering causes lines 20A and 20B in FIG. 3 to merge into a single line 20 separated from line 18 by a vertical displacement equal to V.sub.MB as depicted in FIG. 5b. V.sub.MA equals V.sub.BE + V.sub.SAT again. V.sub.MB may be as low as V.sub.BE + V.sub.SAT. Consequently, V.sub.PSO may again be equal to 1.4 volts.
As the above discussion indicates, the lowest tolerable value for V.sub.PS in the differential amplifier of FIG. 1 is approximately 1.4 volts. In practice, V.sub.PS may have to be a few tenths of a volt higher. While this is still relatively good, it would be quite useful to get down to 1.0 volt or less. For example, a single-cell battery having an unloaded rating of 1.5 volts typically drops down to about 1.0 volt under loaded conditions near the end of its life. The amplifier of FIG. 1 cannot operate efficiently from such a battery.